Prior to description of the prior art, description is made for a technological background concerning the present invention.
FIG. 14 shows a model of a channel with intersymbol interference (ISI) therein.
The model expresses a channel with a finite impulse response (FIR) filter. In the model, a received signal is a synthesized signal synthesized from a lead signal with the output thereof directly received and a delay signal with the output thereof received with a delay due to its reflection or so.
In the figure, a time difference between delay signals is given by a delay circuit DELAY comprising L-segmented shift register. The lead signal is obtained by multiplying a transmitted signal In by channel impulse response (CIR) c0, n as a tap coefficient with a multiplier MULT0. Herein, a subscript n of CIR c0, n indicates a time of data received during TDMA communications.
Also, delay signals are obtained by multiplying delayed transmitted signals In−1 to In−l by tap coefficients c1, n to CL, n with multipliers MULT1 to L respectively. Then, outputs of delay signals from the multipliers MULT0 to L are summed up by a summing device SUM, and an adder (ADD) adds noise Wn to the summed wave outputted from the summing device (SUM) to output the added wave as a received signal rn.
When intersymbol interference (ISI) is not present in the channel, the received signal rn is expressed with the following equation.rn=c0,nIn+Wn  (1) 
In this case, c0, n is a known value, so that a transmitted signal In can easily be estimated from a received signal rn on condition that a noise Wn is small.
By the way, according to the model in FIG. 14, when a transmitted sequence of {In} is transmitted to the channel, this transmitted sequence undergoes intersymbol interference (ISI) in addition to additive white Gaussian noise Wn in the channel. Accordingly, the received signal rn includes not only a time n but also a transmitting sequence In in the past before that time. The received signal at this time is expressed with the following equation:rn=Σci,nIn−i+Wn  (2) wherein the sum Σ is obtained for values of i=0, . . . , L, and L indicates a time length affected by intersymbol interference (ISI), namely a channel memory length.
In the model of the channel shown in FIG. 14, the transmitted sequence In includes a range from time n to time (n−L). An equalizer is often used for the channel described above as a device for estimating a transmitted sequence In from a received signal rn.
Also, when there is frequency offset Δω generated due to a difference between a local oscillator of a transmitter and a local oscillator of a receiver, a received signal is expressed with the following equation:rn=Σci, nIn−i exp(Δωn+θ0)+W′n  (3) wherein θ0 is an initial phase, and W′n is expressed with the following equation:W′n=Wn exp(Δωn+θ0)  (4) 
As described above, the performance of a receiver is deteriorated due to distortion caused by frequency offset Δω in addition to intersymbol interference (ISI). And for this reason, the receiver needs to correct the intersymbol interference (ISI) and also the distortion caused by frequency offset Δω.
Next description is made for an example of a receiver with a frequency offset correcting function based on the conventional technology.
FIG. 15 is a block diagram showing the conventional type of receiver for correcting frequency offset. The receiver in this example is the same as that described in “Method and Device for Compensating Carrier Frequency Offset in TDMA Communication System (Japanese Patent Laid-open Publication No. HEI 6-508244)” disclosed by Lin Yuphan et al.
In FIG. 15, designated at the reference numeral 211 is a CIR estimating circuit for estimating CIR according to a training sequence in a received signal, at 212 a phase deviation detecting circuit for computing a phase deviation according to the CIR estimated value estimated by the CIR estimating circuit 211 and a tail bit described later of the received signal, at 213 an averaging circuit for averaging the phase deviations outputted from the phase deviation detecting circuit 212 and computing a frequency offset estimated value, at 214 a frequency offset correcting circuit for correcting the received signal rn according to the frequency offset estimated value outputted from the averaging circuit 213, and at 215 an equalizer for equalizing the received signal r′n corrected by the frequency offset correcting circuit 214 according to the CIR estimated value outputted from the CIR estimating circuit 211, and estimating the transmitted data sequence.
FIG. 16 shows a burst B1 of received signals received during TDMA communications based on the conventional technology shown in FIG. 15.
In the figure, this burst B1 comprises a training sequence B11, data sequence B12, B13, and tail bits B14, B15, and the training sequence B11 and the tail bits B14, B15 are known in the receiver side.
Next description is made for operations in the example based on the conventional technology with reference to FIG. 15 and FIG. 16.
At first, the CIR estimating circuit 211 computes, when having received a received signal rn, CIR estimated values g0, g1, . . . , gL according to the training sequence B11 in the received burst B1 as shown in FIG. 16 as well as to the training sequence having previously been known in the receiver side.
Then, the phase deviation detecting circuit 212 first computes a phase deviation φm with the following equation according to the CIR estimated values g0, g1, . . . , gL estimated with the known training sequence in the received burst B1 by the CIR estimating circuit 211 as well as to the known tail bits In−L, In−L+1, . . . , In. It should be noted that a subscript m in the equation indicates a phase deviation in m-th received burst.Sn=ΣgiIn−i  (5) φm={Im[rn]·Re[sn]−Im[sn]·Re[rn]}/{ABS[rn]·ABS[sn]}  (6) wherein the sum Σ is obtained for i=0, . . . , L. It should be noted that L indicates, as shown in the channel model in FIG. 14, a time length affected by intersymbol interference (ISI), namely a channel memory length, and corresponds to the number of stages in the delay circuit DELAY. Also, in the equation, designated at the reference sign sn is a replica (estimated value) of a received signal, at Re[a] a real part of a complex number a, at Im[a] an imaginary part of the complex number a, and at ABS[a] an absolute value of the complex number a respectively.
Further, the phase deviation detecting circuit 212 computes a phase deviation per symbol Δφm through the following equation according to the phase deviation φm as described above, and outputs a result of the computing to the averaging circuit 213.Δφm=φm{2/(M−1)}  (7) wherein M indicates a total number of symbols of a received burst B1.
Then, the averaging circuit 213 averages the phase deviation per symbol Δφm estimated for each burst B1, and outputs a result of averaging to the frequency offset correcting circuit 214 as a frequency-offset estimated value Δωm.
The frequency offset correcting circuit 214 corrects frequency offset of a received signal rn through the following equation according to the frequency-offset estimated value Δωm.r′n=rn exp(−jΔωmn)  (8) 
The equalizer 215 estimates transmitted data sequence according to the received signal r′n having been subjected to offset correction outputted from the frequency offset correcting circuit 214 as well as to the CIR estimated value outputted from the CIR estimating circuit 211, and outputs a result of the decision as a decision value.
However, in the receiver with the conventional type of frequency offset correcting function, known data such as tail bits other than the training sequence is required to compute a frequency-offset estimated value, and also a length of tail bits in a received signal is generally required to be longer than a memory length L of the channel, so that transmission efficiency is worse in turn by the length required for the tail bits.
In the example based on the conventional technology, a phase deviation is computed according to the CIR estimated value, the tail bits and the received signal, so that the phase deviation to be detected largely varies with noises. Accordingly, in order to estimate frequency offset with sufficient precision, it is required to suppress variation by making a time constant larger for averaging phase deviations, and for this reason, when frequency offset varies with time, it is difficult to follow the variation in the method described above.
Further, as diversity reception is not performed in the example based on the conventional technology, an error rate in decision is higher as compared to the case where diversity reception is performed.
The present invention has been made for solving the problems as described above, and it is an object of the present invention to provide a receiver with a frequency offset correcting function which has improved the capabilities of being excellent in transmission efficiency without requiring known data other than a training sequence, precisely estimating time-varying frequency offset, and further enabling performance of diversity reception and determination of data at a low error rate.